Electric drive and method for controlling it

ABSTRACT

An electric drive ( 1 ) comprises: a permanent magnet brushless motor ( 2 ), a motor ( 2 ) power supply bridge ( 3 ), a circuit for controlling the power supply bridge ( 3 ) according to rotor position and phase currents (I S ); the drive ( 1 ) comprises a circuit ( 6 ) for detecting the zero crossings of the induced counter electromotive force (E S ) in the stator windings to determine the position of the rotor and a circuit ( 25 ) for indirectly detecting the amplitudes of the phase currents (I S ).

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is the National Phase of International ApplicationPCT/IB2008/002372 filed Sep. 9, 2008 which designated the U.S. and thatInternational Application was published under PCT Article 21(2) inEnglish.

This application claims priority to Italian Patent Application No.BO2007A000619 filed Sep. 12, 2007, and PCT Application No.PCT/IB2008/002382 filed Sep. 9, 2008, which applications areincorporated by reference herein.

TECHNICAL FIELD

This invention relates to an electric drive for a brushless motor withpermanent magnets and to a method for controlling the drive.

By way of non-limiting example, this specification describes a drivecomprising a three-phase brushless motor with permanent magnets thatgenerates a sine-wave counter electromotive force (c.e.m.f.) for drivingaxial, radial and other types of fans used in electric ventilators.

BACKGROUND ART

Considering that the field of application of these electric fans is thatof climate control and cooling systems for installation in motorvehicles, it should be observed that the main aims of developingelectric fans for this purpose are: low acoustic noise, limited energyconsumption and reduced costs.

These requirements have led to the adoption of sine-wave c.e.m.f.brushless motors (AC brushless motors) driven by inverter capable ofgenerating sine-wave currents and making obsolete the use of PWMsix-step driven trapezoidal c.e.m.f. motors (more commonly known as DCbrushless motors).

The sine waveform of the c.e.m.f. and of the related phase currentminimizes active torque ripple (virtually zero), thus reducingmechanical vibrations and acoustic noise.

It is also known that it is possible to minimize current draw togenerate a certain drive torque, thereby maximizing electromechanicalconversion efficiency through optimum drive of AC brushless motors whichare normally driven by current-controlled, impressed voltage inverters.

To obtain this type of drive, the static switches must change state insuch a way that the polar axis of the rotor magnetic field remains at 90electrical degrees to the polar axis of the magnetic field generated bythe current circulating in the stator windings, whatever the torquesupplied and the rotation speed.

To obtain information about the angular position of the rotor,relatively expensive devices are normally used, including absoluteencoders or Hall effect sensors, integral with the stator and suitablypositioned angularly, to detect the sine waveform of the magneticenergizing field along the periphery of the rotor.

The output signals generated by the sensors are then suitably decoded todrive the static switches in such a way as to keep the angular shift of90 electrical degrees between the rotor and stator magnetic fields.

This type of drive requires the use of the above mentioned positionsensors, whose cost is relatively high.

In an attempt to reduce the cost of drives, driving strategies that donot use sensors of this type have been developed.

These driving strategies are based on the consideration that if drive isoptimum, the c.e.m.f. and the phase current are in phase and vice versaat each point in the operating field (torque, rotation speed, DC supplyvoltage).

Consequently, these driving strategies and drives, which have come to beknown as “sensorless”, are based on the reading of electrical quantities(e.g. voltage at motor terminals or current circulating in motorwindings) to detect the points where the c.e.m.f. and the current crosszero (zero crossings), calculate the relative phase between c.e.m.f. andcurrent and implement appropriate methods of driving the inverter staticswitches which tend to keep the two quantities in phase.

One disadvantage of these methods lies in the fact that to detect thezero crossing of the c.e.m.f., that is to say, to read the sign of thec.e.m.f., the current flowing through the windings must remain zero longenough to enable the reading to be taken, which contrasts with thedesired sinusoidal waveform of the current.

For the deviation from the ideal to have negligible effects, the lengthof the time interval during which the current remains zero must bereduced to the minimum and, to eliminate the distortion induced by thecontrolled phase current interruption, however brief, and the risks ofnot reading the desired signal, sophisticated algorithms are introducedto calculate the angular position of the rotor in real time: inpractice, these algorithms are an integral part of field-orientedcontrols (FOC in the jargon of the trade) and require the use ofsophisticated and expensive controllers with high processing capacity(known as DSP controllers in the jargon of the trade).

DISCLOSURE OF THE INVENTION

In this context, the main purpose of the present invention is to proposean electric drive which is free of the above mentioned disadvantages.

One aim of this invention is to provide a low-noise and low energyconsumption drive.

Another aim of the invention is to provide an electric drive based on asimple and inexpensive control architecture.

The stated technical purpose and specified aims are substantiallyachieved by an electric drive with the characteristics described inclaim 1 and in one or more of the dependent claims. The invention alsorelates to a method of controlling the drive.

BRIEF DESCRIPTION OF THE DRAWINGS

Further characteristics and advantages of the invention are moreapparent in the description below, with reference to a preferred,non-limiting embodiment of an electric drive for permanent magnetbrushless motors, as illustrated in the accompanying drawings, in which:

FIG. 1 shows a principle diagram of the electric brushless motor driveaccording to the invention;

FIG. 2 illustrates an equivalent circuit of a phase of an AC brushlessmotor;

FIG. 3 illustrates a vector diagram of the circuit of FIG. 2;

FIG. 4 illustrates a vector diagram representing optimum operation ofthe circuit of FIG. 2;

FIG. 5 is a diagram showing an example of a portion of the driveaccording to the invention;

FIG. 6 shows a circuit diagram of a first detail of the drive of FIG. 1;

FIG. 7 shows a circuit diagram of a second detail of the drive of FIG.1;

FIG. 8 illustrates a procedure for controlling the drive of FIG. 1 untiloptimum operating conditions are reached;

FIG. 9 is a diagram showing the voltages applied to the brushless motor;

FIG. 10 shows the diagram of FIG. 9 in a particular operating condition.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS OF THE INVENTION

With reference to the accompanying drawings and in particular withreference to FIG. 1, the numeral 1 denotes an electric drive accordingto this invention.

This invention is based on the principle of obtaining information,continuous in time, from which to derive the supply voltage values foroptimizing control of the motor powered by the electric drive.

The drive 1 comprises an electric motor 2, for example for driving a fannot illustrated.

As becomes clearer as this description continues, the purpose of thedrive 1 is to obtain information relating to the position of the motor 2rotor by detecting the zero crossing of the counter electromotive force(also referred to in the abbreviated form c.e.m.f.) in a simple andeconomical manner.

By way of example, without limiting the scope of the invention, thisspecification refers to a permanent magnet brushless motor 2 withisotropic, two-pole rotor and three-phase stator winding.

The stator winding comprises three windings with identical shape andnumber of turns, spatially phase-shifted by 120° and connected by a wyeconnection whose centre is not accessible.

FIG. 2 illustrates a circuit model of the motor 2.

Each of the three windings is characterized by a phase resistance R_(S)and a synchronous inductance L_(S).

A voltage V_(S) is applied to the motor 2, while a vector E_(S)represents the induced c.e.m.f. in each of the three stator windings.

The c.e.m.f. has a sine waveform and is due to the rotation of thepermanent magnet rotor; I_(S) is the phase current, also sinusoidal,which flows through each of the three windings.

FIG. 3 shows the vector diagram of the electrical quantities V_(S),I_(S), E_(S) just mentioned.

The direct axis d is oriented in the rotor flow direction φ_(r) and thequadrature axis q makes an angle of 90° with the direct axis d.

According to the law of induction (e=dφ/dt) the induced c.e.m.f. E_(S)in the stator winding is always directed along the quadrature axis q,i.e. it is phase-shifted by 90° with respect to the rotor flow φ_(r).

The voltage V_(S) applied by the drive 1 to the stator windings for apredetermined operating condition is, as mentioned, represented by thevector V_(S).

The stator current vector I_(S) makes an angle Ψ with the vectordifference V_(S)−E_(S). The angle Ψ depends on the characteristicparameters of the motor and on the supply frequency according to therelation:

Ψ=arctan(ωL _(S) /R _(S)).

The electromagnetic power yield of the motor is given by3E_(S)I_(S)cos(γ) where γ is the angle made by E_(S) and I_(S).

The power absorbed by the motor 2 is essentially the sum of theelectromagnetic power yield and of the power losses due to the Jouleeffect in the three phase resistors.

Hence, given a certain electromagnetic yield, the absorbed power islowest when the angle γ is zero, that is, when the c.e.m.f. E_(S) andthe current I_(S) are in phase, as illustrated in FIG. 4.

As illustrated in FIG. 1, the drive 1 comprises a three-phase bridge 3or inverter for powering the motor 2.

Preferably, the drive 1 comprises a low-inductance shunt 3 a connectedas shown in FIG. 1 to the three legs of the inverter 3 and crossed bythe currents circulating in the inverter, as described in more detailbelow.

The drive 1 also comprises a direct current stage 4 for powering thebridge 3 and in turn comprising a levelling capacitor 5 (Cbus) and afiltering inductor 5 a (Lbus).

By way of example, the three-phase bridge 3 generates, throughsine-delta PWM modulation of substantially known type, three voltagesphase-shifted by 120° from each other at variable frequency “freq”.

Advantageously, the amplitude of the fundamental supply voltages can beprogrammed both as a linear function of the frequency f andindependently of the latter.

It should be noticed that, as is well known, the permanent magnetbrushless motor 2 develops torque only at its synchronous speed andtherefore it will rotate exactly at a speed directly proportional to thefrequency f of the applied voltages according to the known relationRPM=120×freq/p, where p is the number of poles of the permanent magnetrotor.

The drive 1 comprises a circuit 6 for detecting the counterelectromotive force and, more specifically, the zero crossings of thec.e.m.f. E_(S), hereinafter also referred to as c.e.m.f. zero crossingdetection circuit 6.

The detection circuit 6 comprises a first stage, illustrated in FIG. 5,and a second stage, illustrated in FIG. 6. The second stage processesthe output signal from the first stage.

In the preferred embodiment illustrated, as will become clearer as thisdescription continues, the information on the position of the rotor isobtained by detecting the zero crossings of the c.e.m.f. generated byonly one of the three phases of the motor.

In alternative embodiments, for example in more sophisticatedapplications that require higher transient response speeds, a rotorposition signal is detected for all the phases by replicating thecircuit described above for each phase.

Considering, for convenience of description, the phase quantities of awye connected motor 2 (it is also known that a delta connected motor isfunctionally indistinguishable from its wye connected equivalent), thevalue of the c.e.m.f. is given by the relation:

$e_{s} = {v_{s} - ( {{R_{s}i_{s}} + {L_{s}\frac{i_{s}}{t}}} )}$

To find the value of e, it is therefore necessary to know both the valuev_(S) of the voltage V_(S) applied to the phase of the motor 2 and theresistive-inductive drop due to the flow of current in the windings ofthe motor 2.

As described below, the drive 1 according to the invention has for anaim to find the resistive-inductive drop and the voltage applied to thephase of the motor 2.

For finding the resistive-inductive drop, the drive 1 comprises, asillustrated in FIG. 5, an inductive-resistive element 9 connected inseries with one of the three phases of the motor 2, as described below.

The element 9 comprises a first inductor L_(i1) and a second inductorL_(i2) with a mutual magnetic coupling coefficient very close to 1.

The two inductors L_(i1) and L_(i2) are connected to form anautotransformer 9 a and, preferably, are wound around a magnetic coreillustrated schematically and labelled 30 in FIG. 5.

By way of example, the magnetic core is in the shape of a double E andis made either of high-frequency ferrite or of plain steel for magneticplates.

The first inductor L_(i1) is connected in series with one of the phasewindings of the motor 2 and constitutes the primary of theautotransformer 9 a.

Preferably, the inductor L_(i1) has a low number N1 of large diameterturns to minimize the power loss due to the Joule effect.

R_(i1) represents the resistance of the winding of the first inductorL_(i1).

The second inductor L_(i2), which constitutes the secondary of theautotransformer, has a number N2 of turns much higher than the number N1of turns of the inductor L_(i1) and is not crossed by the current i_(S),and therefore provides a voltage V_(t2) that depends on the derivativeof the current i_(S) flowing on the primary.

With reference to FIG. 5, if:

V_(t)=voltage at the terminal 10, 11 of FIG. 5, that is, at theterminals of the inductive-resistive element 9;

V_(R)=voltage drop on the resistor R_(i1)

V_(t1)=voltage drop on the first inductor L_(i1);

V_(t2)=voltage drop on the second inductor L_(i2);

M=mutual inductance between L_(i1) and L_(i2);

then:

V_(t) = V_(R) + V_(t 1) + V_(t 2)$V_{t} = {{R_{i\; 1}i_{s}} + {L_{i\; 1}\frac{i_{s}}{t}} + {M\frac{i_{s}}{t}}}$$V_{t} = {{R_{i\; 1}i_{s}} + {( {L_{i\; 1} + M} )\frac{i_{s}}{t}}}$

It is important to note that the expression for V_(t) is formallyidentical to that for the resistive-inductive drop in the windings ofthe motor 2 due to the flow of current.

The following equation can therefore be written:

${{R_{i\; 1}i_{s}} + {( {L_{i\; 1} + M} )\frac{i_{s}}{t}}} = {\alpha ( {{R_{s}i_{s}} + {L_{s}\frac{i_{s}}{t}}} )}$

where

$\alpha = \frac{R_{i\; 1}}{R_{s}}$

is an attenuation coefficient.

Since the synchronous inductance L_(S) of the motor 2, theself-inductance of the primary L_(i1) and the related number of turns N1can be used to find the number of turns N2 of the secondary L_(i2) whichsatisfies the equation:

${N\; 2} = {N\; 1( {\frac{\alpha \; L_{S}}{L_{i\; 1}} - 1} )}$

Thus, by making a resistive-inductive element 9 with the parametersspecified above, the attenuated value of the resistive-inductive drop onthe phase of the motor 2 can be obtained using, in practice, a measuringcircuit corresponding to the equivalent circuit of the phase of themotor 2.

It should be noticed that the attenuation coefficient α indicates theimpact of the c.e.m.f. detection circuit 6 on the total loss of thedrive 1: the lower the coefficient, the lower the loss.

The information on the voltage V_(S) applied to the motor 2 is obtainedusing a circuit 12 for measuring the applied voltage.

The measuring circuit 12 comprises three wye connected resistors 13, 14,15 illustrated in particular in FIG. 5.

By attenuating by the coefficient α both the contribution of the appliedequivalent voltage v_(S) measurable, as described in more detail below,by the set of wye connected resistors 13, 14, 15, and the contributionof the resistive-inductive drop supplied by the mutually coupledinductors L_(i1) and L_(i2), the circuit of FIG. 5 supplies anattenuated c.e.m.f. signal whose amplitude is given by:

${\alpha \cdot e_{S}} = {{\alpha \cdot v_{S}} - {\alpha \cdot R_{S} \cdot i_{S}} - {\alpha \cdot L_{S} \cdot \frac{i_{S}}{t}}}$

Based on the attenuation coefficient α defined above, the wye connectedresistors 13, 14, 15 used for measuring the supply voltage V_(S) issuitably unbalanced.

FIG. 9 shows the real first harmonic voltages V1, V2, V3 generated bythe inverter 3 and applied to the motor 2, each schematicallyrepresented with a respective ideal voltage generator.

As illustrated, the voltage αV1 must appear at the terminals of theresistor 13, also labelled Rα.

Preferably, the values of the resistors 13, 14, 15 can be calculated, ata given point in time, with reference to the symmetrical three-phasetriad that powers the motor 2.

For example, at the point in time where V1 reaches its maximum value Vmthe values of V2 and V3 are −Vm/2.

The circuit to be analysed is therefore the one shown in FIG. 10.

Applying the overlapping effects principle to calculating the voltagedrop on Rα in the circuit of FIG. 10 gives:

$V_{\alpha} = {{V_{m}\frac{R_{\alpha}}{\frac{R}{2} + R_{\alpha}}} + {V_{m}\frac{R_{\alpha} \parallel R}{R_{\alpha} \parallel {R + R}}}}$

from which the relation between Rα and R can be derived for thepreviously fixed value of α:

$R_{\alpha} = {R\frac{\alpha}{3 - {2\alpha}}}$

where R is a generic resistance value for the resistors 14 and 15.

Applying the signal αe_(S) to the signal conditioning circuit shown inFIG. 6 gives a signal for the zero crossing of the counter electromotiveforce that can be processed by a microcontroller 26.

It should be noticed that the behaviour obtained is substantially thesame as that obtained with a digital output Hall sensor.

As illustrated in particular in FIG. 6, this circuit comprises twocomparator stages connected in cascade: the first stage 7, with arespective comparator 16, does not exhibit any hysteresis and spuriousswitching can be detected at its output.

At low rotation speeds, the circuit of FIG. 5 for detecting the zerocrossing of the c.e.m.f. gives voltage values of a few hundredmillivolts and, owing to the low attenuation factor α, the signal-noiseratio is low and causes the above mentioned spurious switching at theoutput of the first stage 7 of the circuit of FIG. 6.

The second stage 8 comprises a second, hysteresis comparator 17 havingan RC input filter 18 to limit the signal oscillations that mighterroneously trigger the comparator 17.

The RC filter 18 comprises a capacitor 19 and a resistor 20 and is ofsubstantially known type.

The second stage 8 also comprises a resistive network to fix theswitching threshold and the related hysteresis.

In the embodiment illustrated, the resistive network comprises fourresistors 21, 22, 23, 24 suitably connected to each other.

Thus, there is no spurious switching at the output of the hysteresiscomparator stage, with obvious advantages in terms of the processingefficiency of the microcontroller 26.

To maximize the efficiency of the motor 2 the current flow in the statorwindings must be in phase with the related c.e.m.f.

With reference to the vector diagram of FIG. 4, it is possible to obtainan approximated expression of the optimum advance angle δ_(opt) for theapplied voltage V_(S) with respect to the c.e.m.f. E_(S).

If the resistive drop in the phase is negligible (the higher theefficiency of the motor 2, the more negligible the drop), assuming K_(E)as the c.e.m.f. constant measured in V/rpm and p as the number of poles,then:

${{{tg}\; \delta_{opt}} \cong \frac{\omega_{el} \cdot {Ls} \cdot {Is}}{Es}} = {{\omega_{el} \cdot {Ls} \cdot {Is} \cdot \frac{\pi \cdot p}{60 \cdot K_{E} \cdot \omega_{el}}} = {\frac{\pi \cdot {Ls} \cdot p}{60 \cdot K_{E}} \cdot I_{S}}}$

Moreover, if the optimum value of the angle δ_(opt) is less than 20electrical degrees, the tangent of the angle can be approximated withthe angle itself and hence:

$\delta_{opt} \cong {\frac{\pi \cdot {Ls} \cdot p}{60 \cdot K_{E}} \cdot I_{S}}$

where the synchronous inductance L_(S) is preferably expressed inHenrys.

In other terms, if the resistive drop R_(S)I_(S) is negligible withrespect to E_(S) and the tangent of the advance angle δ_(opt) can beapproximated with the angle itself, then, in practice, the advance angleδ_(opt) depends linearly only on the phase current I_(S).

Since the drive 1 according to the invention comprises an extremelysimple and economical microcontroller 26, for example an 8-bitmicrocontroller, the above mentioned linear relation between δ_(opt) andthe current draw I_(S); can be stored in it, for example in tableformat, making available to the microcontroller 26 a signal proportionalto the current I_(S) and the microcontroller 26 will be able to controlthe bridge 3 according to the corresponding δ_(opt).

It is for this purpose that the drive 1 comprises a circuit 25 forindirectly detecting phase current amplitudes.

More specifically, the circuit 25 comprises an enveloping detector ordetector stage 27 which processes the voltage signal present at theterminals of the shunt 3 a, directly proportional to the current flowthrough the shunt 3 a itself.

As is known from the literature, the maximum value of the voltage peakson the shunt 3 a is proportional to the phase current peak of the motor2.

Since the phase current is sinusoidal, the reading of the envelopingdetector 27 is equal to the effective value Is of the phase currentmultiplied by √{square root over (2)}.

The enveloping detector 27 keeps track of this information and themicrocontroller 26 samples it at a much lower frequency than that of thecarrier PWM: the validity of the information is guaranteed by the factthat the changing speed of the enveloping detector 27 output, which isdirectly linked to the changing speed of the mechanical load, is verylow.

It should be noticed that the discharge constant of the detector 27 issuitably dimensioned to correctly follow the envelope of the currentpeaks on the shunt 3 a.

Basically, the microcontroller 26, by sampling the output signal of theenveloping detector stage 27 through its analog-to-digital converter,indirectly measures the current value of the phase current and, as aresult, determines the corresponding optimal angle to be applied to keepthe current in phase with the c.e.m.f.

FIG. 7 shows a diagram of an embodiment of the shunt current envelopingdetector 27.

The detector 27 comprises an RC filter 28 for filtering the shuntcurrent envelope.

The detector 27 also comprises a circuit 29 for charging a capacitor insuch a way that when the non-inverting signal is lower than theinverting signal, the capacitor can be discharged through the resistors30 and 31.

The resistors 30 and 31 are suitably connected to enable the device tofollow the shunt current peaks.

FIG. 8 illustrates the procedure for controlling the brushless motor 2,comprising the following steps:

A) step parking or alignment;

B) step of accelerating according to a predetermined ramp V/f;

C) step of “engaging” the c.e.m.f. zero crossing signal, where“engaging” means reaching an optimum operating condition;

D) step of optimized drive.

In steps A and B the inverter 3 powers the motor 2 entirely in “openloop” mode, that is to say, without using either of the two availablefeedback signals, namely, c.e.m.f. zero crossing and shunt currentenvelope.

In step C only the c.e.m.f. zero crossing signal is used.

Lastly, in step D both the c.e.m.f. zero crossing signal and the shuntcurrent envelope signals are used and the inverter 3 drives the motor 2under optimum operating conditions, that is to say, with the c.e.m.f.and the phase current in phase with each other.

During step A three constant voltages are applied to the motor, suitablydefined to enable the current to flow in such a way as to make the rotorturn until it is at a known position where the stator field and therotor field are aligned.

This step ensures that the maximum drive torque possible under “openloop” control conditions can be generated in the next step B.

In step B the motor 2 is powered by three sinusoidal voltages phaseshifted by 120° from each other so as to create a rotary stator field atincreasing frequency and whose amplitude is proportional to thefrequency itself.

During this step, the amplitude of the mean voltage applied to the motor2 varies proportionally with its frequency “freq”, as illustrated inFIG. 8.

The frequency “freq” starts at zero and is increased until it reachesthe “fset” value shown FIG. 8, set in the software of themicrocontroller 26.

The “fset” value is greater than the minimum electrical frequency atwhich the c.e.m.f. zero crossing signal can be surely detected, so thatthe phase relation between the counter electromotive force E_(S) and theapplied voltage V_(S) can be measured in the next steps C and D.

This relation is the angle δ made by the quantities E_(S) and V_(S)shown in FIGS. 3 and 4.

The slope of this acceleration ramp is a parameter of the drive and mustbe modified according to the inertia of the motor 2+load system.

The brushless motor 2 subjected to the rotary stator field generated bythe drive accelerates until it exactly reaches the synchronous speedrelated to “fset”.

In step B the brushless motor 2 is controlled in exactly the same way asan asynchronous motor but, unlike the latter, it reaches the end of rampspeed since, during the ramp itself, the angle between rotor field andstator field never exceeds 90 degrees representing the conditionnecessary and sufficient to generate the drive torque for the permanentmagnet brushless motor which, as is known, is a “synchronous” motor.

The slope value of the ramp V/f is chosen in such a way as to guaranteethat the motor receives sufficient current, and hence torque, toaccelerate it in the required time to the speed corresponding to thefrequency “fset”, for example as a function of environmental parameterssuch as inverter 3 supply voltage and ambient temperature.

Reaching the frequency “fset” triggers step C, during which thefrequency remains constant at the value “fset” and the applied voltageV_(S) is decreased at a predetermined rate.

As mentioned, in step C the c.e.m.f. zero crossing signal is availableand the phase between V_(S) and E_(S) is therefore measured through themicrocontroller 26.

The gradual decrease of V_(S) reduces the current draw I_(S) until itreaches the minimum value required to keep the motor turning: when thiscondition is reached, V_(S) and E_(S) are substantially in phase, themicrocontroller 26 detects the in phase condition between V_(S) andE_(S) and considers step C ended.

Step C is followed by step D.

In step D, only the amplitude Vs is set and not the frequency “freq”.

The microcontroller 26 continuously detects the electrical frequency,acquiring the time interval between two consecutive signal edges,whether homologous or non-homologous, of the output signal from thec.e.m.f. zero crossing detection circuit 6 to which the frequency of theoutput voltage fundamental of the inverter 3 corresponds.

In order to obtain optimum operation of the motor 2, an iterativeprocedure comprising the steps described below is also implemented.

The peak value of the phase current is measured by the microcontroller26 through the shunt current enveloping detector 27.

The microcontroller 26 detects the c.e.m.f. zero crossing through therespective detection circuit 6.

The microcontroller 26 then applies the advance δ_(opt) between V_(S)and E_(S), since the software installed in the microcontroller 26incorporates the relation between the advance angle made by V_(S) andE_(S) and the peak value of the phase current corresponding to optimumoperation.

At this point, the procedure restarts from the measurement of the peakvalue of the phase current.

The above mentioned optimizing procedure causes the brushless motor 2 tooperate with the c.e.m.f. in phase with the respective phase current.

In this situation, as mentioned, absorbed power is minimized; bysuitably setting the time interval of the optimizing procedure, it ispossible to make the system reasonably reactive even to sudden loadvariations due, for example, to temporary choking of the delivery and/orsuction ducts of the air-hydraulic circuit and the subsequent removal ofthe choking itself.

The control method described also allows maximum efficiency of motordrive by causing the current in each stator winding to be in phase withthe respective c.e.m.f.

Thus, the motor generates the maximum torque possible. Expressed inother terms, the brushless motor is driven efficiently because thestator current has no components in the axis d but only in the axis q.

The drive operates in such a way that, once the starting transient isover, motor power consumption is minimized under all load conditions andat all speeds of rotation: in terms of the vector diagram, the phasecurrent is in phase with the respective c.e.m.f.

1. An electric drive comprising: a brushless motor (2) comprising astator and a rotor; a motor (2) power supply bridge (3); means fordetecting the angular position of the rotor; means for detecting thephase current flows in the motor (2); a circuit for controlling thepower supply bridge (3) according to rotor position and phase currents;the drive being characterized in that the means for detecting theangular position of the rotor comprise a circuit (6) for detecting thezero crossings of the counter electromotive force (E_(S)) induced in thestator windings by the rotation of the rotor and in that the means fordetecting the phase currents comprise a circuit (25) for indirectlydetecting the phase currents by enveloping the latter.
 2. The driveaccording to claim 1, characterized in that the circuit (6) fordetecting the zero crossings of the counter electromotive force (E_(S))comprises an inductive-resistive element (9) connected in series with arespective phase of the motor (2) to determine the resistive-inductivedrop due to the current flow in said phase of the motor (2).
 3. Thedrive according to claim 1, characterized in that the circuit (6) fordetecting the zero crossings of the counter electromotive force (E_(S))comprises an inductive-resistive element (9) connected in series witheach phase of the motor (2) to determine the resistive-inductive dropdue to the current flow in each winding of the motor (2).
 4. The driveaccording to claim 2, characterized in that the inductive-resistiveelement (9) comprises a first and a second inductor (L_(i1), L_(i2)),mutually coupled, the first inductor (L_(i1)) in particular beingcrossed by the phase current (I_(S)).
 5. The drive according to claim 4,characterized in that the first and second inductors (L_(i1), L_(i2))are mutually connected to form an autotransformer (9 a), said first andsecond inductors (L_(i1), L_(i2)) being in particular wound around amagnetic core (30).
 6. The drive according to claim 4, characterized inthat the second inductor (L_(i2)) has a number (N2) of turns much higherthan the number (N1) of turns of the first inductor (L_(i1)).
 7. Thedrive according to claim 4, characterized in that the number (N2) ofturns of the second inductor (L_(i2)) is the product of the number (N1)of turns of the first inductor (L_(i1)) by the difference between theratio of the synchronous inductance (L_(S)) of the motor (2) multipliedby an attenuation coefficient (α) and the inductance value (L_(i1)) ofthe first inductor (L_(i1)) and one, that is:${N\; 2} = {N\; {1 \cdot {( {\frac{\alpha \cdot L_{S}}{L_{i\; 1}} - 1} ).}}}$8. The drive according to claim 7, characterized in that the attenuationcoefficient (α) is calculated as the ratio of the resistance (R_(i1)) ofthe first inductor (L_(i1)) to the phase resistance (R_(S)) of the motor(2), that is: $\alpha = {\frac{R_{i\; 1}}{R_{S}}.}$
 9. The driveaccording to claim 1, characterized in that the control circuitcomprises a controller (26) in communication with the circuit (6) thatdetects the zero crossings of the counter electromotive force and withthe circuit (25) that indirectly detects the phase currents (I_(S)), thecontroller (26) controlling the power supply bridge (3) according to thezero crossings of the electromotive force and according to the outputvoltage from the circuit (25) that indirectly detects the phase currents(I_(S)).
 10. The drive according to claim 9, characterized in that itcomprises, stored in the controller (26), the curve of the advance angle(δ) between the supply voltage (V_(S)) and the counter electromotiveforce (E_(S)) as a function of the phase current (I_(S)); said curvebeing in particular linear and obtained in the controller (26) byapproximating tangent of the advance angle (tgδ) with the advance angle(δ) itself.
 11. The drive according to claim 1, characterized in thatthe circuit (6) for detecting the zero crossings of the counterelectromotive force (E_(S)) comprises a first stage (7) in turncomprising a first comparator (16).
 12. The drive according to claim 11,characterized in that the circuit (6) for detecting the zero crossingsof the counter electromotive force (E_(S)) comprises a second stage (8)connected in cascade with the first stage (7) to minimize spuriousswitching.
 13. The drive according to claim 12, characterized in thatthe second stage (8) comprises an hysteresis comparator (17).
 14. Thedrive according to claim 1, characterized in that the circuit (25) thatindirectly detects the phase currents (I_(S)) comprises a shunt resistor(3 a) at the output of the power supply bridge (3) from which theenvelope if the phase currents (I_(S)) can be detected; the circuit (25)that indirectly detects the phase currents (I_(S)) comprising anenveloping detector (27) of the current flow on the shunt resistor (3a).
 15. A method for controlling an electric drive having thecharacteristics described in claim 1, the method being characterized inthat it comprises the steps of: starting the motor (2) by controllingthe power supply bridge (3) in open loop mode, that is to say, withoutusing the signals from the circuit (6) that detects the zero crossingsof the counter electromotive force (E_(S)) and from the circuit (25)that indirectly detects the phase currents; accelerating the motor (2)according to a predetermined ratio between supply voltage and supplyfrequency; decreasing the supply voltage until it reaches the minimumphase current draw value required to keep the motor (2) turning:optimizing the drive by setting only the amplitude of the supplyvoltage.
 16. The method according to claim 15, characterized in thatduring the starting step, three constant voltages are applied to themotor (2), the motor (2) being in particular a three-phase motor, so asto make the rotor turn until it is at a known position.
 17. The methodaccording to claim 16, characterized in that at the known position therotor field and the stator field are aligned.
 18. The method accordingto claim 16, characterized in that it comprises the step of powering themotor (2), in particular after applying the constant voltages, withthree sinusoidal voltages phase shifted by 120° electrical degrees fromeach other so as to create a rotary stator field at increasing frequencywhose amplitude varies linearly with the frequency itself; the statorwinding comprising in particular three windings with identical shape andnumber of turns, spatially phase-shifted by 120° and connected by a wyeconnection whose centre is not accessible.
 19. The method according toclaim 15, characterized in that during the step of accelerating themotor (2) according to a predetermined ratio between supply voltage(V_(S)) and supply frequency, the amplitude of the average voltageapplied to the motor varies linearly with the supply frequency (freq).20. The method according to claim 19, characterized in that the supplyfrequency (freq) starts at zero and is increased until it reaches apredetermined frequency (fset), being in particular a value set in thecontroller (26).
 21. The method according to claim 19, characterized inthat the predetermined frequency (fset) is greater than the minimumelectrical frequency at which the signal of the counter electromotiveforce (E_(S)) zero crossing can be detected by the circuit (6) thatdetects the zero crossings of the counter electromotive force (E_(S)).22. The method according to claim 15, characterized in that during theoptimizing step, the controller (26), by setting only the amplitude ofthe supply voltage, continuously detects the electrical frequency(freq), acquiring the time interval between two consecutive signaledges, whether homologous or non-homologous, of the output signal fromthe circuit (6) that detects the zero crossings of the counterelectromotive force (E_(S)) to which the frequency of the output voltagefundamental of the inverter (3) corresponds.
 23. The method according toclaim 15, characterized in that it comprises an iterative procedure forobtaining the optimized operation of the drive, said procedurecomprising the steps of: measuring the peak value of the phase current(I_(S)) using a shunt current enveloping detector (27); detecting thezero crossings of the counter electromotive force (E_(S)) using thecircuit (6) that detects the zero crossings of the counter electromotiveforce (E_(S)); applying an advance angle (δ) between the supply voltage(V_(S)) and the induced counter electromotive force (E_(S)), inparticular as a function of a relation between the advance angle (δ) andthe peak value of the phase current (I_(S)).
 24. The drive according toclaim 23, characterized in that the predetermined relation sets theadvance angle (δ) by calculating the ratio of the product of pi (π) bythe synchronous inductance (L_(S)) of the motor (2) and the number ofpoles (p) to the product of 60 by counter electromotive force constant(K_(E)), said ratio being further multiplied by the effective value ofthe phase current (I_(S)), that is:$\delta = {\frac{\pi \cdot L_{S} \cdot p}{60 \cdot K_{E}} \cdot I_{S}}$the counter electromotive force constant (K_(E)) being in particularmeasured in volts divided by the number of revolutions per minute, thatis, $\frac{V}{rpm},$ the synchronous inductance (Ls) being expressedHenrys and the phase current (Is) in amps.